Detailed Modelling of a TA7122AP Equivalent Circuit

Detailed Modelling of a
TA7122AP/ECG1085
Equivalent Circuit

• Biasing of Q1 is achieved by linking Pin 5 to Pin 2, typically 100KΩ.
• Voltage-Series Feedback and hence accurate gain is achieved by
  linking a feedback network between Pin 6 and Pin 3.

TA7122/ECG1085 Re-drawn

The TA7122AP or ECG1085 Voltage Amplifier

• For this analysis, Voltage-Series feedback (in blue) is not added to the circuit, but is designed to be employed later to generate an accurate voltage gain at Vo.
  
  
• To the left of Vin is the signal generator, the signal generator's voltage source is Vs, or for this article 𝚫Vo
  
  
• Some circuit component values differ slightly from the original specification, this will be considered in calculations.
  
  
• Parasitic capacitances will be ignored since we are only concerned with audio frequencies.

 

As promised in my last article, a more detailed analysis of the Toshiba TA7122 voltage amplifier now follows. This is not intended to be an electronics tutorial, but more of an attempt to model this old and interesting amplifier, and compare the analysis with actual measured results. Bear in mind both theoretical and actual measurements will have some inherent errors - the latter especially so since readings will be taken from an oscilloscope screen. Nevertheless, a general comparison can be made between theoretical and (near) actual.

I cannot guarantee that the derivation of an expression here is free of errors, but I am reasonably confident of much of the work. Some simplifications were added at the T3 side of the amplifier, where no reference is made to hie, ri, and hfe on that final stage.

It is important to represent the T1 input side of the amplifier accurately, since this is where the bulk of the gain is generated.

I may offer later, a detailed explanation of my thinking as I attempted to mathematically model the TA7122AP IC.

So then, in summary - the purpose of the work below is to establish the voltage gain of the said amplifier as it stands here; that is without Voltage-Series negative feedback that would be later inserted between Vo and the emitter of T1.

The Derivation of Voltage Gain

I will use the delta symbol '𝚫' to highlight what we seek to find.

A prediction of voltage gain 𝚫Vo/𝚫Vs without Voltage-Series feedback now follows.

Much of this is written from an electrical and electronic engineering perspective, and so assumptions of reader knowledge is presumed.

There will be no proofs of Kirchoff's voltage and current laws as this too lengthy to write out. 

Hand-written 'scribbles' and small alterations are included, but this is because this was a 'first run' at collecting all the ideas together. Word processing the work below on a computer would be fraught with difficulties as many readers will know - circuit drawing, finding the correct symbols, lengthy equation writing etc.

 

G1 is the voltage gain for stage 1
G3 could have been written for stage 3, but wasn't.

Setting RL1 = 470KΩ; any higher and the gain
would cause 𝚫Vo to saturate too easily.
With my limited equipment, it can be
difficult to generate a very low input signal 𝚫Vs which
is stable at around 2mV and less.

 

Feedback Network: Above I 'referred' the feedback circuit RF into the input side with an equivalent Thevenin circuit - as if it were all sourced at the input. This then became just one equivalent signal source (but mathematically mixed) with an equivalent input resistance.

 

No additional Thevenin circuit was referred to the T3 side in parallel with Re3, simply because it would have very little effect on calculations. This holds true as Re3 << Rf.

Below should read 'So far then, EQ 6 yields ...'

• K1 and K2 were dimensionless definitions
  that made transposition of formulae
  and computation easier.

Labeling 𝚫VRL2 as 𝚫Vo we have

Matching Theory and Practice
 

• Signal Generator Test Frequencies: 50Hz, 100Hz or 1,000Hz.
• Vcc = 30 volts.
• 𝚫Vs was adjusted so that 𝚫VR2 or 𝚫Vo did not saturate, saturation began a little higher than 25v peak to peak.
• Signal source 𝚫Vs was measured 'offline', ie not connected to the input during 𝚫Vo measurements. Thus effectively creating an accurately measured voltage source.
• Rs is the resistance of the source, and was measured at around 600Ω.
  
  The theoretical model above and its computations were performed on a Spreadsheet, with measured comparisons also included, we can finally compare.
  

Click on this image to see the results.

Increasing Amplifier Gain

To obtain easy control over an effective minute input signal,
a simple potential divider circuit was set up.

A claimed '90dB' gain is quoted for this amplifier if limits of its ability are exploited. With that I decided to push the gain of this near equivalent circuit to as high as I could measure, at least for now. Feedback was reduced, to accomplish this RF was increased from 100KΩ to 1MΩ! This effectively reduces feedback current into the emitter circuit of T1.

 

I needed to set up an additional circuit at the input to create a controllable, but minute driving voltage with my old equipment.

 

With Re1 remaining at 0.22KΩ, and utilizing my old (but calibrated) Hameg 20Mhz oscilloscope to measure the outcome, output was pushed to 10v peak-to-peak, and an effective but new input Vs was measured at a little under 0.25*0.00465 peak-to-peak or 1.1625v p2p. The resultant measured gain was then approximately 20·LOG(10/0.0011625) or 78.7dB.

So how did the mathematical modelling compare? With Rs now being effectively replaced by a Thevenin 0.468KΩ, the spreadsheet computation predicted a gain 9348 with RF = 1MΩ, this equates to 20·LOG(9348) or 79.4dB.

Circuit Issues

 

The amplifier circuit wasn't without problems - with the addition of a 1MΩ feedback resistance, the amplifier's 'switch on' time was delayed, and more so if RF was increased further. 

 

The reason being the base-emitter bias voltage at transistor T1 took more time to reach the Vbe 'cut in' voltage at approximately 0.55v - 0.60volts. What creates this scenario is an effective increased Rf⨉Cbe charging capacitor time constant at the input to T1. With Rf at 1MΩ and transistor base-emitter junction capacitance Cbe (at say ~10pF), we have an increased time constant by a factor of 10. This of course means it will take about ten times as long for T1 to switch on. There may be other factors too why 'switch on' was delayed?

 

To overcome this, use a lower feedback value - recommended 100KΩ or less.

Comment

 

Predicted voltage gain modelling results with actual were good to excellent.

 

Analysis shows that altering hfe, ri (hie), and hfe within the developed equations does not effect the outcomes significantly, and indeed this was verified by the spreadsheet program and so possible variations can largely be ignored.

 

However, dynamic output resistance (1/hoe) or ro can and does effect results. Variations in Re1 also had a strong impact on gain as expected, since this also forms part of a internal current-series negative feedback loop.

 

Regardless of the pleasing accuracy of the mathematical model, ultimately the voltage gain of the TA7122AP/ECG1085 (and their emulation above) will be determined solely by the feedback network from Vo (pin 6) to the emitter of T1 (pin 3). 

 

For official configuration suggestions, see the ECG1085 datasheet online.

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Note: Corrections and modifications to the above article may be periodically made without notice.

29/12/2023

cassettedeckman@gmail.com