Modelling of the TA7122/ECG1085

Modelling of the 1970s
TA7122/ECG1085
Three Stage Voltage Amplifier

Actual circuit is to the right of Vin.

Left hand side represents a signal generator.

The Toshiba TP7122 (and ECG1085?) voltage amplifier was commonly used by Sony and others in their cassette decks and reel to reel machines of the 1970s. It was likely used in many other applications. It also resembles Sanyo’s own LA3122.

The TA7122AP or ECG1085 may indeed be difficult to source today, which is why I decided to investigate building an equivalent, effective configurable voltage amplifier.

The above amplifier comprises of three stages:

• Stage 1 T1 – a high gain voltage amplifier,
• Stage 2 T2 – a buffer, unity gain emitter-follower, and finally
• Stage 3 T3 – a final high gain (but less then Stage 1) voltage amplifier.

The pins as labelled by ECG Semiconductors form part of the powering, and user defined network and feedback system.

Current Shunt feedback from pins 5 to pin 2 help stabilize the amplifier’s gain. Additionally, both T1 and T3 circuits are also configured in current-series feedback topologies. The feedback values of Re in both T1 and T3 stages are small, although I suspect all three Re resistances in each of the emitters are mainly employed to prevent thermal runaway?

To further stabilise the amplifier and thus set a usable gain and bandwidth, voltage-series feedback network from pin 6 to pin 3 is required; marked in blue.


Analysis

Circuit analysis can often be difficult, and so the model offered here is slightly compromised. That is – it is assumed that the operation of this amplifier is restricted to audio frequencies, and no analysis has been made regarding the effects of parasitic capacitances and inductances, resulting in possible, but unlikely self oscillation. Often we see small ceramic capacitors of several pF placed strategically to avoid full phase inversion feeding back to the input – hence self oscillation. 

Stage1 T1: Very High Gain Voltage Amplification

With reference to the circuit above, if we consider just the first stage, and without any feedback from pin 5 to pin 2 (but independently biased), we observe a single stage amplifier with current-series feedback in the emitter leg.

Neglecting parasitic capacitances the voltage gain for Stage 1 can be approximated with reference to the (ac only) small signal h-parameter model ...

Simplified Hybrid Parameter Equivalent
Modelling Only ac Signals.

hie = dynamic input resistance, labelled as ri
1/hoe = dynamic output resistance, labelled as 'ro' below.

Considering the effects of hie, and on the application of Kirchoff's voltage and current laws we arrive at the following Vo vs Vin model -

or alternatively, if the effects of ri and Re1 are ignored (in the left side of the denominator) a simplified model will be ...

In a low loaded state, we can expect the gain to be very high.

Observing the ECG1085 data sheet, several examples are given. With R_L1 commonly set to R_L1=820KΩ, R_E1=0.220KΩ, T1 output dynamic resistance (usually labelled as 1/h_oe, now labelled here as ‘ro’) could be anywhere between 20KΩ and 300KΩ, and so we can expect the gain to be of the order -

(a) ro = 20KΩ

Av = 820*20/(820+20)*(1/0.22) = 88

(b) ro = 300KΩ

Av = 820*300/(820+300)*(1/0.22) = 998

I expect the first stage transistor to be a sensitive high current gain (high h_fe) type, similar to that of the modern and popular KSC1845-FTA.  With such a dramatically variable voltage gain, there will be consequences – the main being saturation and bandwidth. However, more negative feedback will desensitise the amplifier to a more stable manageable system as we will see later.

 

Stage 1: T1 Circuit in Isolation

 

 

This circuit was replicated as indicated in the main figure above, and so was independently configured and biased, with the input test frequency set to just 50Hz.

For any amplifier there is always a gain × bandwidth constant. Since the gain was anticipated to be so high at this stage, clearly the bandwidth was going to be severely compromised. Even at just 500Hz, the gain of this single stage was dropping, so it was decided to set the testing at 50Hz and under, where gain was close to its maximum.

Without presenting biasing data here, setting up the circuit I have chosen RL1=470KΩ (not 820KΩ), and with RE1=0.22KΩ (as before), the voltage gain for this single stage was determined experimentally.

Observing traces on a 10MΩ-configured-input oscilloscope as accurately as possible, I arbitrarily set the input and got: V_RL1=-1.45v, and V_in=2.75mV (not V_S), the computed gain was - 

Av = -1.45v/0.00275 = -527

Again, compare to the theory where Av = -RL1*ro/(RL1+ro)*(1/Re1), then for several estimations of ro, 

 

• ro=50KΩ, Av ~ 205 

• ro=100KΩ, Av ~ 375 

• ro=150KΩ, Av ~ 517 

• ro=200KΩ, Av ~ 638 

 

Then under these biasing conditions, ro (1/hoe) was estimated at around 150kΩ.

Stages 1,2,3: Full Circuit Without External Feedback Network

This time, Vs is measured in an open-circuit state to avoid internal resistances (R_S ~600Ω) interfering with the results. Again, using the oscilloscope visually to compute A_V,

V_S = 7.4mV, and V_RE3 = 20.8v

(peak-to-peak measurements on both accounts)

gives

Av = 2810

This compares with a theoretical approximation of

A_V(CL) = (R_F/R_E)*(R_L3/R_S) = 3833.

(The expression above was derived by modelling the circuit where R_E1=0. Modelling with R_E1 >0 is more complex, I may later write an article on how both were derived!)

Of interest, R_E1 was later short-circuited, and so the small amount of feedback offered by R_E1 was theoretically zeroed, here the gain was examined: V_S = 7.2mV, and V_RE3 = 22v, and so

Av = 3056

A comparison of errors based on the mathematical model were the theoretical was taken as the reference, then for R_E1=0.22KΩ, and R_E1=0KΩ.

R_E1=0.22KΩ, error = |3833-2810|/3833 ~ 26.7%

R_E1=0Ω, error = |3833-3056|/3833 ~ 20.3%.

So, as a stand-alone amplifier, in this present form we can expect a large gain of somewhere between 2500 and 4000, but still with a restricted bandwidth. However, further negative feedback will desensitise the amplifier, and impresses gain and bandwidth stability.

Full Circuit with Final Feedback Network β

According to feedback theory,

A_VCL = A_OL/(1+A_OL·β),
(A_VCL: closed loop gain,  A_OL :open loop gain)

and if A_OL·β>>1 then

A_VCL ~ 1/β,

where β is a feedback ratio (β≡V_E1/Vo) and can be shown via potential divider action to be 

β = R_E1/(R_F2+R_E1)

Applying this to the T1/T2/T3 full circuit -
 

A_VCL= 1/β = (R_F2+R_E1)/R_E1

To drive the amplifier into a stable amplification environment, and much broader bandwidth, we can arbitrarily set R_F2 = 10KΩ, and with R_E1=0.22KΩ,

A_VCL= (10+0.22)/0.22 = 46.5

Comparing this to actual measurement, V_O = 4v (p2p) with V_S = 91.6mV (p2p), we have -

A_VCL = 4/0.0916 = 43.7

Square Wave Response

As a quick indication of the amplifier's stability, the input was subjected square wave excitation at both 1Khz, and 10Khz with a 10KΩ load across Vo. 

Prior to this, I did notice that the amplifier had a very good frequency response, but there was an obvious, but small rise in output at around 100Khz-200Khz. Was this the signal generator?

Recalling amplifier expression A_OL/(1+A_OL·β) again, a rise in output can be attributed to the denominator exhibiting some pronounced phase shifting at high frequencies. The gradual but apparent higher frequency phase shifting suggests that the term |1+A_OL·β| is behaving as |1-A_OL·β|, and thus lowering its value to increase overall gain. Should this amplifier have become very unstable (very unlikely), then we'd presume that the denominator term move closer to zero, ie |1-A_OL·β| → 0.
 
Shown below are traces of two square wave input signals - both at 10Khz. Note the small overshoot on the first, and then compare to the next when a compensating capacitor Cf is added.

 

Please note - The oscilloscope probe leads were compensation calibrated. Indeed, some of this overshoot can be found at the output of the signal generator - possibly a little internal line-capacitance/inductance, and part-signal reflection from generator to input of amplifier, ie impedance mismatch? (NB: this point has not been clarified yet)

On excitation of 10Khz input.
Observe a small amount of overshoot; the amplifier is relatively stable.

With the addition of a compensating 68pF ceramic capacitor across RL2 (Cf), the amplifier is further stabilised, or dampened. This inclusion of Cf is very effective as this revised feedback attenuates the 'lift' around 100Khz-200Khz.

Very stable amplification,
but possibly slightly over-dampened?

In the present setup, the amplifier with a Avcl gain of approximately 44, is delivering a -3dB-0dB-3dB bandwidth which is slightly better than 5Hz - 200Khz.

Amplifier Input and Output Impedance

Not yet measured, but according to feedback theory - for voltage series feedback, Zin increases by a factor of (1+A_OL·β), and Zout by a factor of 1/(1+A_OL·β), ie, a decrease.

Amplifier Distortion

Not yet measured.

*This article is subject to corrections and additions without notice. 18/11/2023

 

cassettedeckman@gmail.com

AKAI AT-2400 Stereo Tuner

 AKAI AT-2400 Stereo Tuner

 

I could not resist this on ebay - an AKAI AT-2400 Stereo Tuner which obviously needed some attention. It was reasonably priced and in a condition which I believed I could turn around.

The marks you see on the fascia are apparently nicotine stains. I certainly took some time to examine all the photographs in the advert for any signs that the tuner had been abused. Apart from the nicotine stains, there are other marks, the most prominent is to the bottom right corner where the brushed aluminum has been subject to some form of impact.

I decided to buy the tuner and on arrival I determined  if the tuner was 'basically' working, which it was. The next task was to disassembled the tuner, and in particular attempt to remove the nicotine from the front.

Servicing Work Undertaken

1. Cleaned fascia, all control knobs and buttons. 
2. Switch-cleaned all contacts and potentiometers, including calibration potentiometers.
3. Replaced all electrolytic capacitors.
4. Calibrated FM and AM where possible without the use of an oscilloscope (≥150Mhz bandwidth required) and RF signal generator.
5. Adjusted the 'High Blend' mixing control to reduced the perception of stereo white/pink noise for weaker, or noisy stereo FM signals. 
6. Replaced all Power Supply Unit (PSU) high voltage rated protection-capacitors for the rectifying diodes.
   
7. Replaced PSU rectifying diodes.
8. Replaced the 13v zener diode in the PSU.
   

Pending Work: Replace the PSU voltage regulation main transistor.

Shown below is a temporary replacement bridge rectifier and new high voltage diode-protection capacitors that protect the rectifying diodes - mainly during switch-on. At switch-on, there is often a high current surge or spike where the diodes are subjected to a high rate of change of voltage (high dv/dt) and thus high current. At all times, capacitors exhibit the electrical characteristic of i=CᐧdV/dt amperes, and so these will help to bi-pass excessive transient current surge or spike.

After completing the first 9 tasks, the tuner is working fine, and looking good.

View From the Outside

The AKAI AT-2400 sits below the Trio KA-6100.

Adjusting High Blend Stereo Mixing

What is 'High Blend' as defined by AKAI?

From what I can glean from the the service manual, it appears that 'High Blend' is a technique to reduce background high frequency low level interference, or white/pink noise when reception conditions are not favourable.

To achieve this, the stereo image is narrowed with much greater emphasis placed on higher audio frequencies. Background noise is far more noticeable if the noise is stereophonic, rather than monophonic. Monophonic noise is 'in phase', in contrast - stereophonic is not.

The method AKAI employed was to mix left and right channels, but only with increasing audio frequency.

Observing the diagram below partially illustrates how they did this.

The Sanyo LA3122 IC is a dual channel high gain voltage amplifier, similar to a modern OP Amp today. The Voltage-Series negative feedback network loop from pin 5 (or 10) to pin 3 (or 12) dictates gain and applied filtering characteristics if needed.

To the left of this circuit, is the 'High Blend' mixing unit - a simple but effective capacitor circuit of just 15nF is placed across both left and right channel inputs. Since capacitive reactance is given by 1/{2𝛑fC} Ω, increasing either capacitance C, or frequency f will lower the reactance (ie, lower impedance), and so at higher capacitance or higher frequencies the audio is mixed and placed towards the centre of the stereo image.

To make this more effective, the original C14 was swapped for a 22nF polyester capacitor. I may even try 27nF at a later date?

Sanyo LA3155 and LA3122

Sanyo's LA3122 and LA3155 high gain voltage amplifiers have been used for potentially many applications including: audio phono (moving magnet) cartridge amplification with RIAA playback de-emphasis, IEC/NAB de-emphasis for tape recorder playback, an active low pass or high pass filter, and of course as a simple voltage amplifier with constant gain.

The LA3155/3122 series is centered around a three stage NPN voltage amplifier with an open loop gain of several thousand.

With reference to the LA3122, the voltage gain in Open Loop Gain configuration can be approximated to: Av ~ {(220/0.4)*(4.7/0.3)} or around 8600. This figure is likely to be subject to some considerable error, for one reason, there will later be some internal negative feedback in the form of current-shunt from pin 4 to pin 2. And another, no consideration was made of the base input resistances of each transistor.

A more analytical approach at audio frequencies would suggest that the gain will be of the magnitude: Av ~ [{(220*hie)/(220+hie)}*1/0.4] ✕ [(4.7*hie)/(4.7+hie)*1/0.3]. Here two-ported h-parameter hie is the transistor's input resistance under biased conditions, and can vary from 20KΩ to 300KΩ (may be less and more?) depending on the type of the transistor. Stage 1 is a very high gain stage, and typically hie ~ 100KΩ ... 300KΩ, stage2 is just an emitter-follower buffer, and at stage 3, we can expect hie ~ 20KΩ ... 100KΩ. Setting hie1 ~ 300KΩ and hie2 ~ 100KΩ, we arrive at an approximation of 4700; much less than 8600, and possibly more realistic? Such an amplifier is gain variable, and its bandwidth will be very restricted.

Applied negative feedback effectively desensitizes any amplifier, and in this case the current-shunt mentioned above, also concurrently applies bias to the base of the first transistor.

Whatever the default gain is, this becomes almost irrelevant. Given that amplifier voltage gain is very high and potentially variable, applying external negative feedback will finally determine the amplifier's overall voltage gain, broaden its bandwidth considerably, and offer excellent voltage gain stability. Apply external negative feedback and we have a closed loop system.

Closed Loop Gain

With the implementation of external voltage-series feedback, Rf (or Zf if it contains reactive elements) active in circuit, the gain can be shown to be Av/(1+Av*𝛃).

And if Av*𝛃 ≫ 1 (which is often desired), then the Closed Loop Gain will gravitate towards A ~ 1/𝛃. At this point, voltage gain is almost independent of open loop gain Av.

The 'beta' term 𝛃 is defined as the feedback ratio, which in voltage-series feedback topology is 𝛃=Vf/Vo.

In the particular circuit above we have -

𝛃 = Vf/Vo = 400/(400+Zf).

(This ratio is derived from potential divider action)

Simply connecting an impedance Zf (or resistance Rf) between pin 5 and pin 3 will result in a voltage gain of 

A = 1/𝛃
or

A = (400+Zf)/400. 

Sanyo's LA3122 Circuit Diagram:
Sanyo recommend 220kΩ for first stage biasing,
and is also part of internal current-shunt feedback.
Zf is the feedback impedance.

Finally, Sanyo state a maximum gain of 40dB in their datasheet example above at 1,000Hz. This can be verified both experimentally, and from 1/𝛃 or (400+39000)/400 ~ 98.5.

That is, in decibel form ...

20*LOG(98.5) = 20*1.99 ~ 40dB.

This article is subject to possible changes and corrections.
09/11/2023.

cassettedeckman@gmail.com